In the field of external radiotherapy with charged particles, adrotherapy, which uses protons and carbon ions and other ion species, is one of the most advanced therapies, affording a finite penetration depth, low deposition of energy at input and marked fall-off, namely, the distal decay of the distribution of the dose. However, some of its benefits may become a risk for the patient on account of the uncertainties during administration of the treatment. It is hence of fundamental importance to carry out monitoring during the radiotherapeutic treatment and regular calibration of the physical parameters of the beam of particles.
This is typically carried out by installing at output from the particle accelerator ionization chambers (with a low value of water equivalence, typically less than 1 mm) and via Quality Assurance (QA) procedures, which are carried out on a daily basis in health-care facilities equipped with different devices and instruments for each parameter to be verified.
The interaction of proton and ion beams with human tissue (which is mainly made up of water) enables the majority of the dose to be conveyed to a precise depth, following the profile of the so-called Bragg peak. In this way, it is possible to increase the precision on the target, limiting the dose that may reach healthy tissues. Moreover, the distal and lateral fall-off of a proton beam is considerably better than the lateral penumbra caused by a photon beam, enabling a fast decay of the dose in the vicinity of adjacent critical structures. As a consequence, the total energy deposited in a patient for a given target dose is lower than that of conventional treatments using photons. However, the Bragg peak for a single-energy proton and ion beam is so narrow that only a limited interval of depth can be treated with a very high dose. In order to widen the interval of treatment depth and supply a uniform dose on the tumour, a spread-out Bragg peak (SOBP) is created as set of pure peaks sent at a decreasing depth (by varying the energy of the particles) and with a reduced dose to obtain the desired modulation. FIG. 3 shows the dose DO from non-modulated proton beams (pure Bragg peak BP) and modulated proton beams (spread-out Bragg peak SOBP). It also shows the sets of peaks SP, the amplitude of which is weighted. FIG. 3 also indicates the interval EI of deposition of the energy of the spread-out peak.
Since protons and ions deposit their energy dose in a relatively small volume, corresponding to the interval EI, it is of fundamental importance to verify correctly the position of the interval EI of deposition of the beam of particles prior to treatment of patients. Calibration of the instrument (size, shape, and intensity of the beam) and verification of the depth-dose curves are carried out during the QA procedures.
According to the prior art, it is known to use for this purpose small ionization chambers or diodes that move through dummies made of tissue-equivalent materials (for example, water or perspex). In this context, it has been suggested to use multilayer ionization chambers (MLICs) in order to accelerate the QA procedures in health-care facilities, as is described, for example, in Lin, S. et al., (2009) “A multilayer ionization chamber for proton beam Bragg peak curve measurements”, Proceedings of the International Conference of the Particle Therapy Co-Operative Group (PTCOG), Heidelberg.
The principle that superintends use of the above apparatuses is the possibility of measuring the charge deposited on each of the anodes (or cathodes) of the various layers of the device while the beam of particles passes through a stack of calibrated water-equivalent absorbers. This enables instantaneous evaluation of the depth-dose distribution of the beam (whether single-energy beam or spread-out Bragg peaks) by virtue of the simultaneous reading of all the ionization chambers that make up the MLIC device. The typical structure of an MLIC is illustrated in FIG. 1.
FIG. 1 shows in this connection an MLIC sensor 10, which comprises a plurality of sensor structures, or channels, 20, in FIG. 1 201, . . . , 20N, comprising pairs 151, . . . , 15N of anodes 11 and cathodes 12, separated by an ionization chamber 14, i.e., a space that identifies the ionization region, and followed by an absorber layer 13.
The total number of channels identifies the maximum energy range of the particles that can be measured, whereas the materials and the physical thicknesses of each channel determine the water-equivalent thickness of the MLIC sensor 10. Known MLIC sensors comprise a fixed number of channels, for example 128 or 180 channels.
FIG. 2 illustrates an accelerator 1000, which is able to generate a beam of charged particles 1300 conveyed through a tunnel 1100 and an outlet mouth 1200 in the direction of a spatial region P, which in the figure is represented as being cylindrical of a length equal to the interval EI of deposition of the energy of the spread-out peak, even though in general it may assume also other shapes. This spatial region P is located, for example, above a therapy table 1400 on which a patient to be treated can be positioned. In the spatial region P, for purposes of calibration, an apparatus for the calibration of beams of charged particles 100 is positioned in such a way that the beam of particles 1300 passes through MLIC sensors 20 that make up the calibration apparatus 100. This apparatus 100 is modular; i.e., it comprises a plurality of aligned sensor modules.
In FIG. 2 it may be noted how the modular MLIC calibration apparatus 100 comprises an outer casing 110, of a substantially parallelepipedal shape, which comprises within it a horizontal stack of modular elements 120, each including an MLIC sensor 10, or another type of sensor for measuring the position and size of the beam of particles, and a supporting frame for the aforesaid sensor.
The calibration apparatus 100 enables instantaneous evaluation of the characteristics of the therapeutic beam of particles 1300 in the directions X, Y, and Z, where Z corresponds to the direction along which the depth D is evaluated.
In this framework, where the instantaneous flow is very high and the efficiency of the calibration and measurement apparatuses must be controlled and possibly corrected, and moreover, the readout front-end of the detector and of its channels must be able to cover the entire range of expected input signals, particular attention must be dedicated to the circuit for acquisition of the signal from the channels of the detector or sensor.
Generally known are ASIC (Application-Specific Integrated Circuit) circuit solutions based upon a count-type charge converter.
These circuits comprise, in the first place, a plurality of channel circuits. Each channel circuit converts the input charge into an increment of a given number of counts of a purposely provided counter, where a fixed amount of charge, in what follows referred to as “quantum of charge”, corresponds to the count of one.
A schematic representation of a channel circuit 205 is shown in FIG. 4.
An input current i is integrated by means of an integrator circuit 201 comprising an operational transconductance amplifier (OTA) 202, an input resistance Rin connected to the inverting terminal of the OTA 202, and a capacitance Cint between the inverting terminal and the output of the OTA 202. An output voltage Vout of the integrator 201, at output from the OTA 202, increases when the input current i is negative, i.e., it comes out of the channel circuit 205, whereas it decreases otherwise. The output voltage Vout of the integrator 201 is compared with two fixed thresholds, a high one VTH and a low one VTL, via two synchronous comparators CMP_1 and CMP_2, the other input of which is connected to the output voltage Vout of the integrator 201. The threshold voltage of each of the two comparators CMP_1 and CMP_2 is set from outside: the high threshold voltage VTH is the voltage threshold of the first comparator CMP_1 that operates on negative input currents i, whereas the low threshold voltage VTL regards the comparator CMP_2, which is active for positive input currents. The value of the thresholds VTH and VTL does not have a particular influence on operation of the channel circuit 205 as long as the voltage variation remains within the output range of the OTA 202 and as long as the difference VTH-VTL between the two thresholds is greater than the voltage jump caused by subtraction of the charge quantum.
The comparators CMP_1 and CMP_2 have their own outputs connected to two control inputs of a pulse generator PG. Appearing in FIG. 4 is also a periodic clock signal clk, which provides a synchronisation reference both to the comparators CMP_1 and CMP_2 and to the pulse generator PG.
The input node A of the OTA 202 is connected to a reference voltage VR, which is also the node on which the current pulse of polarity opposite to the one used for subtraction of charge is discharged, operating through two switches in series, a first switch sw1 and a second switch sw2.
The reference voltage VR is a voltage that is connected to guard rings in the sensor to ensure that any possible charges that are not directly generated by ionization, but are for example generated by the difference of potential present between layers, are collected by the guard rings, without being transmitted to the layer on which the measurement signal (i.e., the current i) is picked up, which thus collects prevalently just the charges produced by ionization of the gas traversed by the beam of charged particles.
The channel circuit 205 further comprises a charge-control capacitance Csub, one terminal of which is connected to the node identified between the first switch sw and the second switch sw2.
Whenever the input voltage of the comparator CMP_1 or CMP_2, i.e., the output voltage Vout of the integrator 201, crosses the respective threshold VTH or VTL, the corresponding comparator CMP_1 or CMP_2 sets at its output a given logic level for the input of a pulse generator PG. As long as this input level is set, the pulse generator PG sends to a capacitor Csub a positive pulse PC on its other terminal, as shown in FIG. 5. The voltage amplitude of the pulse PC, ΔVpulse, is defined as the difference between two reference voltages Vp+ and Vp−, which are set from outside.
The total capacitance Csub may be obtained with three capacitors in parallel, of 50, 100, and 200 fF respectively, which may be added independently, so that the charge-control capacitance Csub can be selected with each value between 50 and 350 fF in steps of 50 fF, via a capacitance-selection signal Cap_sel. The output response of the capacitance Csub to a voltage pulse PC is represented by two current signals of opposite sign, δ+ and δ−, associated to which are charges Q+ and Q−, respectively, according to the following relation (1):Q+=Csub·ΔVpulseQ−=Csub·(−ΔVpulse)  (1)
The timing of the first current signal δ+ is determined by the rising edge of the pulse PC, whereas the negative current signal δ− corresponds to the falling edge of the pulse.
By operating on the timing of the command pulses P1 and P2 of the switches sw1 and sw2, the charge Q+ or Q− can be directed either to the input of the OTA 202 or else to the reference voltage VR, adding or subtracting a fixed amount of charge at each pulse P1, P2 generated by the pulse generator PG. The decision on which of the charge signals is to be sent to the OTA 202 depends upon the output of the comparators, in order to remove a fixed amount of charge from the capacitor Cint of the integrator. This results in a change in the voltage drop on the capacitor Cint, which is given by Q/Cint.
FIG. 5 shows a timing chart that represents the pulse PC sent to the capacitor Csub, the command pulses P1, P2 of the switches sw1, sw2, and the currents δ+ and δ− in the switches.
If, after the action described above, the input voltage, i.e., the output Vout of the integrator 201 of the comparator CMP_1 (or CMP_2) remains above (below) the threshold VTH (or VTL), the pulse generator PG continues to emit pulses PC and stops when the voltage Vout passes again below (above) the threshold.
In parallel, the pulse generator PG sends a count pulse, which may be a signal for increment Cnt_Up or a signal for decrement Cnt_Down of the up/down counter 220, according to which is the comparator that is acting at its input.
All the operations are synchronized via the external master clock clk and controlled via a digital finite-state machine (FSM) implemented in the generator block PG and not shown in FIG. 4. If at a cycle of the clock signal the pulse generator PG detects that the comparator CMP_1 (or CMP_2) is active, the next two clock cycles are used for generating the pulses for charge subtraction and command of the switches sw1/sw2. Two supplementary clock cycles are required before re-activation of the FSM in the pulse generator PG in order to leave time for the OTA 202 to reduce the output voltage. Thus, to generate a count, five master-clock pulses are required.
The relation between the frequency of counts ν and the input current i is:
                    v        =                  i                      Q            c                                              (        2        )            where Qc is the quantum of charge, which is given by:Qc=Csub·ΔVpulse  (3)
Reading of the total charge collected in the detector is provided by the number of counts generated during the measurement time multiplied by the value of the charge quantum Qc.
FIG. 6 shows a known solution of circuit arrangement, designated by 200, which envisages operating, for example, with a measurement apparatus 100 that has available a number M of channels 20 comprised in one or more MLIC sensors 10 arranged in a horizontal stack, as described with reference to FIG. 2. It is here emphasized how reference to a sensor 10 of an MLIC type is provided purely by way of non-limiting example, it being possible for the sensor to be also of some other ionization-chamber type suitable for detecting the intensity of beams of charged particles and configured for sending its own measurement signal through a plurality of measurement channels 20, for example a pixel ionization-chamber detector. The circuit arrangement 200 comprises a number N equal to 64 of inputs in0, . . . , in63, on each of which a channel 20 can be connected for conveying thereto a corresponding current i collected by the channel 20 of the MLIC sensor 10.
Each of the above inputs in0, . . . , in63 in the circuit arrangement 200 is the input of a respective circuit branch B0, . . . , B63. Each k-th branch Bk comprises a channel conversion circuit 205 including a respective current-to-frequency converter 210, which converts the respective current i at input to the k-th branch into a frequency νk. This frequency νk is then measured by a counter 220, comprised in the k-th branch Bk, which supplies at output the value of the counter CT, in particular a 32-bit value.
The values assumed by the counters CT0, . . . , CT63 of the branches B0, . . . , B63 are supplied as inputs to a multiplexer 250, which, upon command from a 6-bit selection bus CS, enables selection of the channel 20 from the channels connected to the inputs in0, . . . , in63, i.e., of the branch from the branches B0, . . . , B63, to be sent at its own output as output data O and to be acquired via a readout electronics 400 not shown in FIG. 4. It should be noted that, in effect, the current-to-frequency converter implemented by the circuit arrangement 200 corresponds to a charge-to-count converter.
As may be noted, in the circuit arrangement 200 of FIG. 4 also shown at input are the signals, supplied by an external controller, that have the function of controlling the channel circuits 205, hence the thresholds VTH, VTL of the comparators, the voltages VP+, VP−, the capacitance-selection signal Cap_sel, in addition to the signal on the selection bus CS of the chip and the selection circuits of the multiplexer 250. The circuit 200 moreover receives a latch signal, L, which has the function of loading at a given instant the output of the counters 220 into as many registers, and moreover an integrator-reset signal RA and a counter-reset signal RD. The integration capacitor Cint of each channel circuit 205 may in fact be discharged through a common digital reset input RA. Likewise, all the counters 220 may be zeroed via a common reset asynchronous digital input, RD.
The circuit of FIG. 4 is described, for example, in the paper by La Rosa et al., Nuclear Instruments and Methods in Physics Research A 583 (2007) 461-468 and manages, via a single channel branch, a respective channel.
However, in this type of circuit, even using a maximum conversion rate of 20 MHz and configuring the quantum of charge to the maximum value (for example, 1.155 pC), the maximum current that a branch can convert before saturating is less than 24 μA. This value is too low for applications with pulsed accelerators, such as synchrocyclotrons. With these instruments, with a duty cycle of the pulse of a few units per thousand, the instantaneous current during the pulse must reach values greater than those of current applications with linear accelerators, even by two or three orders of magnitude.
Consequently, the known solutions present limits in the values of amplitude of current on the channels that they can manage.